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TM
MP2354
2A, 23V, 380KHz Step-Down Converter
The Future of Analog IC Technology
TM
DESCRIPTION
The MP2354 is a monolithic step down switch mode converter with a built in internal power MOSFET. It achieves 2A continuous output current over a wide input supply range with excellent load and line regulation. Current mode operation provides fast transient response and eases loop stabilization. Fault condition protection includes cycle-by-cycle current limiting and thermal shutdown. In shutdown mode the regulator draws 20A of supply current.
FEATURES
* * * * * * * * * * * * * * * 0.18 Internal Power MOSFET Switch Stable with Low ESR Output Ceramic Capacitors Up to 95% Efficiency 2A Output Current Wide 4.75V to 23V Operating Input Range Fixed 380KHz Frequency Thermal Shutdown Cycle-by-Cycle Over Current Protection Programmable Under Voltage Lockout Frequency Synchronization Input Operating Temperature: -40C to +85C Available in an 8-Pin SO Package Distributed Power Systems Battery Chargers Pre-Regulator for Linear Regulators
EVALUATION BOARD REFERENCE
Board Number EV2354DS-00A Dimensions 2.3"X x 1.4"Y x 0.5"Z
APPLICATIONS
"MPS" and "The Future of Analog IC Technology" are Trademarks of Monolithic Power Systems, Inc.
TYPICAL APPLICATION
INPUT 4.75V to 23V
3 2
Efficiency vs Output Current
10nF
VIN RUN BST LX
95 90
EFFICIENCY (%)
OPEN AUTOMATIC STARTUP OPEN IF NOT USED
5.0V 3.3V 2.5V
8 1
4 6
MP2354
SYNC GND 5 FB COMP 7
B230A
OUTPUT 3.3V / 2A
85 80 75 70 65 60 0 0.5 1.0 1.5
3.3nF
2.0
2.5
MP2354_TAC_S01
OUTPUT CURRENT (A)
MP2354_TAC_EC01
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
PACKAGE REFERENCE
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage (VIN) .................................... 25V Switch Voltage (VLX) ....................... -1V to +26V Bootstrap Voltage (VBST)....................... VLX + 6V Feedback Voltage (VFB) ................... -0.3 to +6V Enable/UVLO Voltage (VRUN)........... -0.3 to +6V Comp Voltage (VCOMP) ..................... -0.3 to +6V Sync Voltage (VSYNC) ....................... -0.3 to +6V Junction Temperature...............................150C Lead Temperature ....................................260C Storage Temperature ..............-65C to +150C
TOP VIEW
SYNC BST VIN LX 1 2 3 4 8 7 6 5 RUN COMP FB GND
Recommended Operating Conditions
(2)
MP2354_PD01-SOIC8
Input Voltage (VIN) ......................... 4.75V to 23V Operating Temperature .............-40C to +85C
Thermal Resistance
Part Number* MP2354DS * Package SOIC8 Temperature -40C to +85C
(3)
SOIC8.................................... 105 ..... 50... C/W
Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1" square of 1 oz copper.
JA
JC
For Tape & Reel, add suffix -Z (eg. MP2354DS-Z) For Lead Free, add suffix -LF (eg. MP2354DS-LF-Z)
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25C, unless otherwise noted.
Parameter Feedback Voltage Symbol Condition VFB 4.75V VIN 23V Min 1.198 Typ 1.222 0.18 10 0 3.4 1.95 400 IC = 10A VFB = 0V Sync Drive 0V to 2.7V VFB = 1.0V VFB = 1.5V 500 342 445 90 0 700 380 35 1000 418 600 Max 1.246 Units V A A A/V V/V A/V KHz KHz KHz % %
Upper Switch On Resistance RDS(ON)1 Lower Switch On Resistance RDS(ON)2 Upper Switch Leakage Current Limit (4) Current Sense Transconductance GCS Output Current to Comp Pin Voltage Error Amplifier Voltage Gain AVEA Error Amplifier GEA Transconductance Oscillator Frequency fS Short Circuit Frequency Sync Frequency Maximum Duty Cycle DMAX Minimum Duty Cycle DMIN
VRUN = 0V, VLX = 0V 2.7
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MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 12V, TA = +25C, unless otherwise noted.
Parameter RUN Shutdown Threshold RUN Pull Up Current EN UVLO Threshold Rising EN UVLO Threshold Hysteresis Supply Current (Shutdown) Supply Current (Quiescent) Thermal Shutdown
Note: 4) Equivalent output current = 1.5A 50% Duty Cycle 2.0A 50% Duty Cycle Assumes ripple current = 30% of load current. Slope compensation changes current limit above 40% duty cycle.
Symbol Condition ICC > 100A VRUN = 0V VEN Rising
Min 0.7 1.0 2.37
Typ 1.0 1.3 2.5 210
Max 1.3 2.62
Units V A V mV
VRUN 0.4V VRUN 2.8V, VFB = 1.5V
20 1.0 155
35 1.2
A mA C
PIN FUNCTIONS
Pin # 1 Name SYNC Description Synchronization Input. This pin is used to synchronize the internal oscillator frequency to an external source. There is an internal 11k pull down resistor to GND, therefore leave SYNC unconnected if unused. Bootstrap (C5). This capacitor is needed to drive the power switch's gate above the supply voltage. It is connected between LX and BST pins to form a floating supply across the power switch driver. The voltage across C5 is about 5V and is supplied by the internal +5V supply when the LX pin voltage is low. Supply Voltage. The MP2354 operates from a +4.75V to +23V unregulated input. C1 is needed to prevent large voltage spikes from appearing at the input. Switch. This connects the inductor to either VIN through M1 or to GND through M2. Ground. This pin is the voltage reference for the regulated output voltage. For this reason care must be taken in its layout. This node should be placed outside of the D1 to C1 ground path to prevent switching current spikes from inducing voltage noise into the part. Feedback. An external resistor divider from the output to GND, tapped to the FB pin sets the output voltage. To prevent current limit run away during a short circuit fault condition the frequency foldback comparator lowers the oscillator frequency when the FB voltage is below 700mV. Compensation. This node is the output of the transconductance error amplifier and the input to the current comparator. Frequency compensation is done at this node by connecting a series R-C to ground. See the Compensation section for exact details. Enable/UVLO. A voltage greater than 2.62V enables operation. Leave RUN unconnected for automatic startup. An Under Voltage Lockout (UVLO) function can be implemented by the addition of a resistor divider from VIN to GND. For complete low current shutdown it's the RUN pin voltage needs to be less than 700mV.
2
BST
3 4 5
VIN LX GND
6
FB
7
COMP
8
RUN
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
Circuit of Figure 2, VIN = 12V, VO = 3.3V, L1 = 15H, C1 = 10F, C2 = 22F, TA = +25C, unless otherwise noted.
Heavy Load Operation
2A Load
VO, AC 50mV/div. VIN, AC 200mV/div. IL 1A/div. VO, AC 20mV/div. VIN, AC 20mV/div. IL 1A/div.
Light Load Operation
No Load
VLX 10V/div.
VLX 10V/div.
MP2354-TPC01
MP2354-TPC02
Startup from Shutdown
2A Resistive Load
Load Transient
VO, AC 200mV/div.
VRUN 2V/div.
VOUT 1V/div.
IL 1A/div. ILOAD 1A/div.
IL 1A/div.
MP2354-TPC03
MP2354-TPC04
Short Circuit Protection
Short Circuit Recovery
VOUT 2V/div.
VOUT 2V/div.
IL 1A/div.
IL 1A/div.
MP2354-TPC05
MP2354-TPC06
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
OPERATION
The MP2354 is a current mode regulator. The COMP pin voltage is proportional to the peak inductor current. At the beginning of a cycle: the upper transistor M1 is off; the lower transistor M2 is on (refer to Figure 1), the COMP pin voltage is higher than the current sense amplifier output; and the current comparator's output is low. The rising edge of the 380KHz CLK signal sets the RS Flip-Flop. Its output turns off M2 and turns on M1 thus connecting the SW pin and inductor to the input supply. The increasing inductor current is sensed and amplified by the Current Sense Amplifier. Ramp compensation is summed to Current Sense Amplifier output and compared to the Error Amplifier output by the Current Comparator. When the Current Sense Amplifier plus Slope Compensation signal exceeds the COMP pin voltage, the RS Flip-Flop is reset and the MP2354 reverts to its initial M1 off, M2 on state. If the Current Sense Amplifier plus Slope Compensation signal does not exceed the COMP voltage, then the falling edge of the CLK resets the Flip-Flop. The output of the Error Amplifier integrates the voltage difference between the feedback and the 1.23V bandgap reference. The polarity is such that an FB pin voltage lower than 1.222V increases the COMP pin voltage. Since the COMP pin voltage is proportional to the peak inductor current an increase in its voltage increases current delivered to the output. The lower 10 switch ensures that the bootstrap capacitor voltage is charged during light load conditions. External Schottky Diode D1 carries the inductor current when M1 is off.
VIN 3 INTERNAL REGULATORS SYNC 1 OSCILLATOR SLOPE COMP CLK CURRENT SENSE AMPLIFIER + -5V
35/380kHz +
2 Q Q 4
BST
+
S R
0.7V RUN 8
--
SHUTDOWN COMPARATOR LOCKOUT COMPARATOR
--
CURRENT COMPARATOR
LX
-2.50V/ 2.29V
1.8V + -5 GND
+
FREQUENCY FOLDBACK COMPARATOR
--
0.7V 6
1.22V FB
+
ERROR AMPLIFIER 7 COMP
MP2354_BD01
Figure 1--Functional Block Diagram
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
INPUT 4.75V to 23V OPEN AUTOMATIC STARTUP OPEN IF NOT USED
8 1
C5 10nF
3 VIN RUN 2 BST LX 4 6
MP2354
SYNC GND 5 FB COMP 7
D1 B230A
OUTPUT 2.5V / 2A
C6 OPEN
C3 3.3nF
MP2354_TAC_F02
Figure 2--Typical Application Circuit Sync Pin Operation Inductor The SYNC pin driving waveform should be a The inductor is required to supply constant square wave with a rise time less than 20ns. current to the output load while being driven by Minimum High voltage level is 2.7V. Low level the switched input voltage. A larger value is less than 0.8V. The frequency of the external inductor will result in less ripple current that will sync signal needs to be greater than 445KHz. result in lower output ripple voltage. However, the larger value inductor will have a larger A rising edge on the SYNC pin forces a reset of physical size, higher series resistance, and/or the oscillator. The upper transistor M1 is lower saturation current. A good rule for switched off immediately if it is not already off. determining the inductance to use is to allow 250ns later M1 turns on connecting LX to VIN. the peak-to-peak ripple current in the inductor Setting the Output Voltage to be approximately 30% of the maximum The output voltage is set using a resistive switch current limit. Also, make sure that the voltage divider from the output to FB (see peak inductor current is below the maximum Figure 2). The voltage divider divides the output switch current limit. The inductance value can voltage down by the ratio: be calculated by:
VFB = VOUT x R2 R1 + R2
L= VOUT V x 1 - OUT fS x IL VIN
Where VFB is the feedback voltage and VOUT is the output voltage. Thus the output voltage is:
VOUT = 1.23 x (R1 + R2) R2
Where VIN is the input voltage, fS is the 380KHz switching frequency, and IL is the peak-topeak inductor ripple current. Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by:
ILP = ILOAD + VOUT V x 1 - OUT 2 x fS x L VIN
R2 can be as high as 100k, but a typical value is 10k. Using that value, R1 is determined by:
R1 = 8.18 x (VOUT - 1.23 )(k )
For example, for a 3.3V output voltage, R2 is 10k, and R1 is 17k.
Where ILOAD is the load current.
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
Table 1 lists a number of suitable inductors from various manufacturers. The choice of which style inductor to use mainly depends on the price vs. size requirements and any EMI requirement. Table 1--Inductor Selection Guide
Package Dimensions (mm) W L H
7.0 7.3 5.5 5.5 6.7 10.1 5.0 7.6 10.0 9.7 9.4 9.4 7.8 8.0 5.7 5.7 6.7 10.0 5.0 7.6 10.0 1.5 13.0 13.0 5.5 5.2 5.5 5.5 3.0 3.0 3.0 5.1 4.3 4.0 3.0 5.1
For simplification, choose the input capacitor whose RMS current rating greater than half of the maximum load current. The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. 0.1F, should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The input voltage ripple caused by capacitance can be estimated by:
VIN = ILOAD V V x OUT x 1 - OUT fS x C1 VIN VIN
Vendor/ Model Sumida
CR75 CDH74 CDRH5D28 CDRH5D28 CDRH6D28 CDRH104R
Core Type
Open Open Shielded Shielded Shielded Shielded Shielded Shielded Shielded Open Open Open
Core Material
Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite Ferrite
Toko
D53LC Type A D75C D104C D10FL
Output Capacitor The output capacitor is required to maintain the DC output voltage. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by:
VOUT = VOUT V x 1 - OUT fS x L VIN 1 x R ESR + 8 x f S x C2
Coilcraft
DO3308 DO3316
Input Capacitor The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice. Choose X5R or X7R dielectrics when using ceramic capacitors. Since the input capacitor (C1) absorbs the input switching current it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by:
I C1 = ILOAD
V V x OUT x1- OUT VIN VIN
Where L is the inductor value, C2 is the output capacitance value and RESR is the equivalent series resistance (ESR) value of the output capacitor. In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance. The output voltage ripple is mainly caused by the capacitance. For simplification, the output voltage ripple can be estimated by:
VOUT = VOUT 8 x fS
2
V x 1 - OUT VIN x L x C2

The worst-case condition occurs where:
I C1 I = LOAD 2
In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to:
VOUT = VOUT V x 1 - OUT x R ESR fS x L VIN
The characteristics of the output capacitor also affect the stability of the regulation system. The
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MP2354 Rev. 1.4 1/6/2006
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MP2354 can be optimized for a wide range of capacitance and ESR values. Output Rectifier Diode The output rectifier diode supplies the current to the inductor when the upper transistor M1 is off. Use a Schottky diode to reduce losses due to the diode forward voltage and recovery times. Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current.
The DC gain of the voltage feedback loop is given by:
A VDC = R LOAD x G CS x A VEA x VFB VOUT
Where AVEA is the error amplifier voltage gain, 400V/V; GCS is the current sense transconductance, 1.95A/V; RLOAD is the load resistor value. The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at:
fP1 = GEA 2 x C3 x A VEA
Table 2 provides the Schottky diode part numbers based on the maximum input voltage and current rating. Table 2--Schottky Rectifier Selection Guide
VIN (Max) 15V 20V 2A Load Current Part Number Vendor (5) 30BQ015 4 B220 1 SK23 6 SR22 6 20BQ030 4 B230 1 SK23 6 SR23 3, 6 SS23 2, 3
Note: 5) Refer to Table 3 for Rectifier Manufacturers
Compensation MP2354 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP pin is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system.
M IN PS D TE CO O R N NA NF O L ID T D US EN IS E TI T R ON AL IB LY U T E
fP2 = 1 2 x C2 x R LOAD
Where GEA is the transconductance, 770A/V.
error
amplifier
The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:
f Z1 = 1 2 x C3 x R3
26V
Table 3--Schottky Diode Manufacturers
Vendor Diodes, Inc. Fairchild Semiconductor General Semiconductor International Rectifier On Semiconductor Pan Jit International
# 1 2 3 4 5 6
Web Site www.diodes.com www.fairchildsemi.com www.gensemi.com www.irf.com www.onsemi.com www.panjit.com.tw
Smaller fZ1 provides more phase margin, but longer transient settling time. A trade-off has to be made between the stability and the transient response. A typical value is less than one-fourth of the crossover frequency. The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at:
fESR = 1 2 x C2 x R ESR
TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
In this case, a third pole set by compensation capacitor (C6) and compensation resistor (R3) is used compensate the effect of the ESR zero on loop gain. This pole is located at:
f P3 = 1 2 x C6 x R3
the the to the
To optimize the compensation components for conditions not listed in Table 4, the following procedure can be used. 1) Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation:
R3 = 2 x C2 x f C VOUT x G EA x G CS VFB
The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is important. Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system unstable. A good rule of thumb is to set the crossover frequency to approximately onetenth of the switching frequency. Switching frequency for the MP2354 is 380KHz, so the desired crossover frequency is around 38KHz. Table 4 lists the typical values of compensation components for some standard output voltages with various output capacitors and inductors. The values of the compensation components have been optimized for fast transient responses and good stability at given conditions. Table 4--Compensation Values for Typical Output Voltage/Capacitor Combinations
VOUT
2.5V 3.3V 5V 12V 2.5V 3.3V 5V 12V
2) Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, fZ1, to less than one forth of the crossover frequency provides sufficient phase margin. Determine the C3 value by the following equation:
C3 > 4 2 x R3 x f C
Where R3 is the compensation resistor value. 3) Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the 380KHz switching frequency, or the following relationship is valid:
f 1 L1
10H min. 15H min. 15H min. 22H min. 10H min. 15H min. 15H min. 22H min.
C2
22F Ceramic 22F Ceramic 22F Ceramic 22F Ceramic 560F Al. 30m ESR 560F Al 30m ESR 470F Al. 30m ESR 220F Al. 30m ESR
R3
5.6k 7.5k 11k 27k 140k 187k 237k 267k
C3
4.7nF 3.3nF 2.2nF 1nF 1nF 1nF 1nF 1nF
C6
None None None None 120pF 82pF 56pF 22pF
If this is the case, then add the second compensation capacitor (C6) to set the pole fP3 at the location of the ESR zero. Determine the C6 value by the equation:
C6 = C2 x RESR R3
MP2354 Rev. 1.4 1/6/2006
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TM
MP2354 - 2A, 23V, 380KHz STEP-DOWN CONVERTER
External Bootstrap Diode It is recommended that an external bootstrap diode be added when the system has a 5V fixed input or the power supply generates a 5V output. This helps improve the efficiency of the regulator. The bootstrap diode can be a low cost one such as IN4148 or BAT54.
5V
BS
MP2354
SW
10nF
MP2354_F03
Figure 3--External Bootstrap Diode This diode is also recommended for high duty cycle operation (when
VOUT >65%) and high VIN
output voltage (VOUT>12V) applications.
PACKAGE INFORMATION
SOIC8
PIN 1 IDENT. 0.229(5.820) 0.244(6.200)
0.150(3.810) 0.157(4.000)
0.0075(0.191) 0.0098(0.249)
SEE DETAIL "A"
0.013(0.330) 0.020(0.508) 0.050(1.270)BSC 0.011(0.280) x 45o 0.020(0.508)
0.189(4.800) 0.197(5.004) 0.053(1.350) 0.068(1.730) 0.049(1.250) 0.060(1.524)
0o-8o
0.016(0.410) 0.050(1.270)
DETAIL "A"
SEATING PLANE 0.001(0.030) 0.004(0.101) NOTE: 1) Control dimension is in inches. Dimension in bracket is millimeters.
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
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